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rev. d information furnished by analog devices is believed to be accurate and reliable. however, no responsibility is assumed by analog devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. no license is granted by implication or otherwise under any patent or patent rights of analog devices. a tmp01 * one technology way, p.o. box 9106, norwood, ma 02062-9106, u.s.a. tel: 781/329-4700 www.analog.com fax: 781/326-8703 ? analog devices, inc., 2002 low power programmable temperature controller functional block diagram vptat v+ temperature sensor and voltage reference 2.5v sensor 1 2 3 4 8 7 6 5 hysteresis generator window comparator r1 tmp01 vref set high set low gnd r2 r3 under over features C558c to +1258c (C678f to +2578f) operation 61.08c accuracy over temperature (typ) temperature-proportional voltage output user-programmable temperature trip points user-programmable hysteresis 20 ma open collector trip point outputs ttl/cmos compatible single-supply operation (4.5 v to 13.2 v) low-cost 8-pin dip and so packages applications over/under temperature sensor and alarm board level temperature sensing temperature controllers electronic thermostats thermal protection hvac systems industrial process control remote sensors general description the tmp01 is a temperature sensor that generates a voltage output proportional to absolute temperature and a control signal from one of two outputs when the device is either above or below a specific temperature range. both the high/low temperature trip points and hysteresis (overshoot) band are determined by user- selected external resistors. for high volume production, these resistors are available on-board. the tmp01 consists of a band gap voltage reference combined with a pair of matched comparators. the reference provides both a constant 2.5 v output and a voltage proportional to absolute temperature (vptat) which has a precise temperature coefficient of 5 mv/k and is 1.49 v (nominal) at 25 c. the comparators compare vptat with the externally set tempera- ture trip points and generate an open-collector output signal when one of their respective thresholds has been exceeded. * protected by u.s. patent no. 5,195,827. hysteresis is also programmed by the external resistor chain and is determined by the total current drawn out of the 2.5 v refer- ence. this current is mirrored and used to generate a hysteresis offset voltage of the appropriate polarity after a comparator has been tripped. the comparators are connected in parallel, which guarantees that there is no hysteresis overlap and eliminates erratic transitions between adjacent trip zones. the tmp01 utilizes proprietary thin-film resistors in conjunc- tion with production laser trimming to maintain a temperature accuracy of 1 c (typical) over the rated temperature range, with excellent linearity. the open-collector outputs are capable of sinking 20 ma, enabling the tmp01 to drive control relays directly. operating from a 5 v supply, quiescent current is only 500 a (max). the tmp01 is available in low-cost 8-pin epoxy mini-dip and so (small outline) packages.
rev. d C2C tmp01fp, tmp01es/tmp01fs?pecifications plastic dip and surface mount packages (v+ = 5 v, gnd = o v, ?0 c t a +85 c, unless otherwise noted.) parameter symbol conditions min typ max unit inputs set high, set low offset voltage v os 0.25 mv offset voltage drift tcv os 3 v/ c input bias current, ? i b 25 50 na input bias current, ? i b 25 100 na output vptat 1 output voltage vptat t a = 25 c, no load 1.49 v scale factor tc vptat 5 mv/k temperature accuracy, ? t a = 25 c, no load ?.5 0.5 1.5 c temperature accuracy, ? t a = 25 c, no load ? 1.0 3 c temperature accuracy, ? 10 c < t a < 40 c, no load 0.75 c temperature accuracy, ? 10 c < t a < 40 c, no load 1.5 c temperature accuracy, ? ?0 c < t a < 85 c, no load ?.0 1 3.0 c temperature accuracy, ? ?0 c < t a < 85 c, no load ?.0 2 5.0 c temperature accuracy, ? ?5 c < t a < 125 c, no load 1.5 c temperature accuracy, ? ?5 c < t a < 125 c, no load 2.5 c repeatability error 4 ? vptat 0.25 degree long-term drift error 2,6 0.25 0.5 degree power supply rejection ratio psrr t a = 25 c, 4.5 v v+ 13.2 v 0.02 0.1 %/v output vref output voltage, ? vref t a = 25 c, no load 2.495 2.500 2.505 v output voltage, ? vref t a = 25 c, no load 2.490 2.500 2.510 v output voltage, ? vref ?0 c < t a < 85 c, no load 2.490 2.500 2.510 v output voltage, ? vref ?0 c < t a < 85 c, no load 2.485 2.500 2.515 v output voltage, ? vref ?5 c < t a < 125 c, no load 2.5 0.01 v output voltage, ? vref ?5 c < t a < 125 c, no load 2.5 0.015 v drift tc vref ?0 ppm/ c line regulation 4.5 v v+ 13.2 v 0.01 0.05 %/v load regulation 10 a i vref 500 a 0.1 0.25 %/ma output current, zero hysteresis i vref 7 a hysteresis current scale factor 1 sf hys 5.0 a/ c turn-on settling time to rated accuracy 25 s open-collector outputs over, under output low voltage v ol i sink = 1.6 ma 0.25 0.4 v v ol i sink = 20 ma 0.6 v output leakage current i oh v+ = 12 v 1 100 a fall time t hl see test load 40 ns power supply supply range v+ 4.5 13.2 v supply current i sy unloaded, +v = 5 v 400 500 a i sy unloaded, +v = 13.2 v 450 800 a power dissipation p diss +v = 5 v 2.0 2.5 mw notes 1 k = c + 273.15. 2 guaranteed but not tested. 3 does not consider errors caused by heating due to dissipation of output load currents. 4 maximum deviation between 25 c readings after temperature cycling between ?5 c and +125 c. 5 typical values indicate performance measured at t a = 25 c. 6 observed in a group sample over an accelerated life test of 500 hours at 150 c. specifications subject to change without notice. test load 20pf 1k v+
rev. d C3C tmp01 tmp01 tmp01fj?pecifications to-99 metal can package (v+ = 5 v, gnd = o v, ?0 c t a +85 c, unless otherwise noted.) parameter symbol conditions min typ max unit inputs set high, set low offset voltage v os 0.25 mv offset voltage drift tcv os 3 v/ c input bias current, ? i b 25 100 na output vptat 1 output voltage vptat t a = 25 c, no load 1.49 v scale factor tc vptat 5 mv/k temperature accuracy, ? t a = 25 c, no load ? 1.0 3 c 10 c < t a < 40 c, no load 1.5 c ?0 c < t a < 85 c, no load ?.0 2 5.0 c ?5 c < t a < 125 c, no load 2.5 c repeatability error 4 ? vptat 0.25 degree long-term drift error 2,6 0.25 0.5 degree power supply rejection ratio psrr t a = 25 c, 4.5 v v+ 13.2 v 0.02 0.1 %/v output vref output voltage, ? vref t a = 25 c, no load 2.490 2.500 2.510 v vref ?0 c < t a < 85 c, no load 2.480 2.500 2.520 v vref ?5 c < t a < 125 c, no load 2.5 0.015 v drift tc vref ?0 ppm/ c line regulation 4.5 v v+ 13.2 v 0.01 0.05 %/v load regulation 10 a i vref 500 a 0.1 0.25 %/ma output current, zero hysteresis i vref 7 a hysteresis current scale factor 1 sf hys 5.0 a/ c turn-on settling time to rated accuracy 25 s open-collector outputs over, under output low voltage v ol i sink = 1.6 ma 0.25 0.4 v v ol i sink = 20 ma 0.6 v output leakage current i oh v+ = 12 v 1 100 a fall time 2 t hl see test load 40 ns power supply supply range v+ 4.5 13.2 v supply current i sy unloaded, +v = 5 v 400 500 a i sy unloaded, +v = 13.2 v 450 800 a power dissipation p diss +v = 5 v 2.0 2.5 mw notes 1 k = c + 273.15. 2 guaranteed but not tested. 3 does not consider errors caused by heating due to dissipation of output load currents. 4 maximum deviation between 25 c readings after temperature cycling between ?5 c and +125 c. 5 typical values indicate performance measured at t a = 25 c. 6 observed in a group sample over an accelerated life test of 500 hours at 150 c. specifications subject to change without notice.
rev. d tmp01 C4C caution esd (electrostatic discharge) sensitive device. electrostatic charges as high as 4000 v readily accumulate on the human body and test equipment and can discharge without detection. although the tmp01 features proprietary esd protection circuitry, permanent damage may occur on devices subjected to high-energy electrostatic discharges. therefore, proper esd precautions are recommended to avoid performance degradation or loss of functionality. warning! esd sensitive device absolute maximum ratings 1 maximum supply voltage . . . . . . . . . . . . . . . . ?.3 v to +15 v maximum input voltage (sethigh, setlow) . . . . . . . . . ?.3 v to [(v+) +0.3 v] maximum output current (vref, vptat) . . . . . . . . . 2 ma maximum output current (open-collector outputs) . . 50 ma maximum output voltage (open-collector outputs) . . . . 15 v operating temperature range . . . . . . . . . . . ?5 c to +150 c dice junction temperature . . . . . . . . . . . . . . . . . . . . . . 150 c storage temperature range . . . . . . . . . . . . 65 c to +150 c lead temperature (soldering 60 sec) . . . . . . . . . . . . . . 300 c notes 1 stresses above those listed under absolute maximum ratings may cause perma- nent damage to the device. this is a stress rating; functional operation at or above this specification is not implied. exposure to the above maximum rating conditions for extended periods may affect device reliability. 2 digital inputs and outputs are protected, however, permanent damage may occur on unprotected units from high energy electrostatic fields. keep units in conduc- tive foam or packaging at all times until ready to use. use proper antistatic handling procedures. 3 remove power before inserting or removing units from their sockets. package type ja jc unit 8-pin plastic dip (p) 103 1 43 c/w 8-lead soic (s) 158 2 43 c/w 8-lead to-99 can (j) 150 1 18 c/w notes 1 ja is specified for device in socket (worst-case conditions). 2 ja is specified for device mounted on pcb. ordering guide temperature package package model/grade range l description option tmp01fp xind plastic dip n-8 tmp01es xind soic so-8 tmp01fs xind soic so-8 tmp01fj 2 xind to-99 can h-08a notes 1 xind = ?0 c to +85 c. 2 consult factory for availability of mil/883 version in to-99 can. general description the tmp01 is a linear voltage-output temperature sensor, with a window comparator that can be programmed by the user to activate one of two open-collector outputs when a predeter- mined temperature setpoint voltage has been exceeded. a low drift voltage reference is available for setpoint programming. the temperature sensor is basically a very accurate, temperature compensated, band gap-type voltage reference with a buffered output voltage proportional to absolute temperature (vptat), accurately trimmed to a scale factor of 5 mv/k. see the applica- tions information following. the low drift 2.5 v reference output vref is easily divided externally with fixed resistors or potentiometers to accurately establish the programmed heat/cool setpoints, independent of temperature. alternatively, the setpoint voltages can be supplied by other ground referenced voltage sources such as user- programmed dacs or controllers. the high and low setpoint voltages are compared to the temperature sensor voltage, thus creating a two-temperature thermostat function. in addition, the total output current of the reference (i vref ) determines the magnitude of the temperature hysteresis band. the open collector outputs of the comparators can be used to control a wide variety of devices. vptat v+ enable tmp01 vref set high set low gnd 7 8 1 2 3 4 6 hysteresis current current mirror i hys voltage reference and sensor 1k hysteresis voltage 5 temperature output window comparator under over figure 2. detailed block diagram
rev. d C5C typical performance characteristics tmp01 20 5 015 10 supply voltage ?v supply current ? a 550 350 400 375 450 425 475 500 525 +25 c +125 c +85 c ?5 c ?0 c tpc 1. supply current vs. supply voltage 5.0 3.0 4.5 3.5 4.0 75 125 50 100 75 50 25 0 25 temperature c minimum supply voltage v tpc 2. minimum supply voltage vs. temperature 75 125 50 100 75 50 25 0 25 +2.0 3.0 +1.0 1.0 0 +1.5 +0.5 0.5 1.5 temperature c vptat error c v+ = 5v tpc 3. vptat accuracy vs. temperature 2.508 2.496 2.500 2.504 2.506 2.502 2.498 75 125 50 100 75 50 25 0 25 temperature c vref v v+ = 5v tpc 4. vref accuracy vs. temperature 6.0 0 3.0 1.0 2.0 5.0 4.0 50 10 040 30 20 v c = 15v v+ = 5v t a = 25 c i c ma v ce v tpc 5. open-collector output ( over , under ) saturation voltage vs. output current 2.510 2.490 2.496 2.492 2.494 2.502 2.498 2.500 2.504 2.506 2.508 1000 200 0 800 400 600 x + 3 x x 3 curves not normalized extrapolated from operating life data t = hours of operation at 125 c; v+ = 5v vref v *+,4&5 *

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rev. d tmp01 C6C 100 1k 1m 100k 10k 20 100 40 20 0 60 80 frequency hz psrr db v+ = 5v i vref = 10 a tpc 7. vref power supply rejection vs. frequency 1.0 0.1 0.01 75 50 125 100 75 50 25 0 25 offset voltage mv v+ = 5v i vref = 7.5 a tpc 8. set high, set low input offset voltage vs. temperature 8 0 2 1 4 3 5 6 7 0.4 0.24 0.32 0 0.08 0.16 0.16 0.08 offset mv number of devices v+ = 5v t a = 25 c i vref = 5 a tpc 9. comparator input offset distribution 7.2 6.2 7 6.8 6.6 6.4 8 7.8 7.6 7.4 reference current a number of devices 10 0 2 1 4 3 5 6 7 8 9 v+ = 5v t a = 25 c tpc 10. zero hysteresis current distribution
rev. d tmp01 C7C temperature hysteresis the temperature hysteresis is the number of degrees beyond the original setpoint t emperature that must be sensed by the tmp01 before the setpoint comparator will be reset and the output disabled. figure 2 shows the hysteresis profile. the hysteresis is programmed by the user by setting a specific load on the refer- ence voltage output vref. this output current i vref is also called the hysteresis current, which is mirrored internally and fed to a buffer with an analog switch. lo hi output voltage over, under temperature hysteresis low hysteresis high = hysteresis low t setlow t sethigh hysteresis high figure 3. tmp01 hysteresis profile after a temperature setpoint has been exceeded and a comparator tripped, the buffer output is enabled. the output is a current of the appropriate polarity which generates a hysteresis offset volt- age across an internal 1000 ? resistor at the comparator input. the comparator output remains ?n?until the voltage at the comparator input, now equal to the temperature sensor voltage vptat summed with the hysteresis offset, has returned to the programmed setpoint voltage. the comparator then returns low, deactivating the open-collector output and disabling the hysteresis current buffer output. the scale factor for the pro- grammed hysteresis current is: i hys = i vref = 5 a / c + 7 a thus since vref = 2.5 v, with a reference load resistance of 357 k ? or greater (output current 7 a or less), the temperature setpoint hysteresis will be zero degrees. see the temperature programming discussion below. larger values of load resistance will only decrease the output current below 7 a and will have no effect on the operation of the device. the amount of hyster- esis is determined by selecting a value of load resistance for vref, as shown below. programming the tmp01 in the basic fixed-setpoint application utilizing a simple resistor ladder voltage divider, the desired temperature setpoints are programmed in the following sequence: 1. select the desired hysteresis temperature. 2. calculate the hysteresis current i vref . 3. select the desired setpoint temperatures. 4. calculate the individual resistor divider ladder values needed to develop the desired comparator setpoint voltages at sethigh and setlow. the hysteresis current is readily calculated, as shown. for example, for 2 degrees of hysteresis, i vref = 17 a. next, the setpoint voltages v sethigh and v setlow are determined using the vptat scale factor of 5 mv/k = 5 mv/( c + 273.15), which is 1.49 v for 25 c. then calculate the divider resistors, based on those setpoints. the equations used to calculate the resistors are: v sethigh = (t sethigh + 273.15)(5 mv / c ) v setlow = (t setlow + 273.15) (5 mv / c ) r 1 ( k ? ) = (v v ref ?v sethigh )/i vref = = (2.5 v ? v sethigh )/i vref r 2 ( k ? ) = (v sethigh ?v setlow )/i vref r 3 ( k ? ) = v setlow /i vref 1 2 3 4 8 7 6 5 (v vref v sethigh )/i vref = r1 tmp01 (v sethigh v setlow )/i vref = r2 v setlow /i vref = r3 v sethigh v setlow v vref = 2.5v i vref gnd v+ vptat under over figure 4. tmp01 setpoint programming the total r1 + r2 + r3 is equal to the load resistance needed to draw the desired hysteresis current from the reference, or i vref . the formulas shown above are also helpful in understanding the calculation of temperature setpoint voltages in circuits other than the standard two-temperature thermostat. if a setpoint function is not needed, the appropriate comparator should be disabled. sethigh can be disabled by tying it to v+, setlow by tying it to gnd. e ither output can be left uncon nected. 218 248 273 298 323 348 373 398 55 25 18 0 25 50 75 100 125 67 25 0 32 50 77 100 150 200 212 257 vptat k c f 1.09 1.24 1.365 1.49 1.615 1.74 1.865 1.99 figure 5. temperaturevptat scale
rev. d tmp01 C8C understanding error sources the accuracy of the vptat sensor output is well characterized and specified; however, preserving this accuracy in a heating or cooling control system requires some attention to minimizing the various potential error sources. the internal sources of setpoint programming error include the initial tolerances and temperature drifts of the reference voltage vref, the setpoint comparator input offset voltage and bias current, and the hyster- esis current scale factor. when evaluating setpoint programming errors, remember that any vref error contribution at the com- parator inputs is reduced by the resistor divider ratios. the comparator input bias current (inputs sethigh, setlow) drops to less than 1 na (typ) when the comparator is tripped. this can account for some setpoint voltage error, equal to the change in bias current times the effective setpoint divider ladder resistance to ground. the thermal mass of the tmp01 package and the degree of ther- mal coupling to the surrounding circuitry are the largest factors in determining the rate of thermal settling, which ultimately determines the rate at which the desired temperature measure- ment accuracy may be reached. thus, allow sufficient time for the device to reach the final temperature. the typical thermal time constant for the plastic package is approximately 140 sec- onds in still air. therefore, to reach the final temperature accuracy within 1%, for a temperature change of 60 degrees, a settling time of 5 time constants, or 12 minutes, is necessary. the setpoint comparator input offset voltage and zero hyster- esis current affect setpoint error. while the 7 a zero hysteresis current allows the user to program the tmp01 with moderate resistor divider values, it does vary somewhat from device to device, causing slight variations in the actual hysteresis obtained in practice. comparator input offset directly impacts the pro- grammed setpoint voltage and thus the resulting hysteresis band, and must be included in error calculations. external error sources to consider are the accuracy of the pro- gramming resistors, grounding error voltages, and the overall problem of thermal gradients. the accuracy of the external programming resistors directly impacts the resulting setpoint accuracy. thus, in fixed-temperature applications, the user should select resistor tolerances appropriate to the desired program- ming accuracy. resistor temperature drift must be taken into account also. this effect can be minimized by selecting good quality components, and by keeping all components in close thermal proximity. applications requiring high measurement accuracy require great attention to detail regarding thermal gradients. careful circuit board layout, component placement, and protection from stray air currents are necessary to minimize common thermal error sources. also, the user should take care to keep the bottom of the setpoint programming divider ladder as close to gnd (pin 4) as possible to minimize errors due to ir voltage drops and coupling of exter- nal noise sources. in any case, a 0.1 f capacitor for power supply bypassing is always recommended at the chip. safety considerations in heating and cooling system design designers should anticipate potential system fault conditions which may result in significant safety hazards which are outside the control of and cannot be corrected by the tmp01-based circuit. governmental and industrial regulations regarding safety requirements and standards for such designs should be observed where applicable. applications information self-heating effects in some applications, the user should consider the effects of self-heating due to the power dissipated by the open-collector outputs, which are capable of sinking 20 ma continuously. under full load, the tm p01 open-collector output device is dissipating p diss = 0.6 v . 020 a = 12 mw which in a surface-mount so package accounts for a tempera- ture increase due to self-heating of ? t = p diss ja = .012 w 158 c/w = 1.9 c. this will, of course, directly affect the accuracy of the tmp01 and will, for example, cause the device to switch the heating output ?ff?2 degrees early. alternatively, bonding the same package to a moderate heatsink limits the self-heating effect to approximately ? t = p diss jc = .012 w 43 c/w = 0.52 c. which is a much more tolerable error in most systems. the vref and vptat outputs are also capable of delivering sufficient current to contribute heating effects and should not be ignored. buffering the voltage reference as mentioned before, the reference output vref is used to generate the temperature setpoint programming voltages for the tmp01 and also is used to determine the hysteresis tempera- ture band by the reference load current i vref . the on-board output buffer amplifier is typically capable of 500 a output drive into as much as 50 pf load (max). exceeding this load will affect the accuracy of the reference voltage, could cause thermal sensing errors due to dissipation, and may induce oscillations. selection of a low drift buffer functioning as a voltage follower with high input impedance will ensure optimal reference accuracy, and will not affect the programmed hysteresis current. amplifi- ers which offer the low drift, low power consumption, and low cost appropriate to this application include the op295, and members of the op90, op97, op177 families, and others as shown in the following applications circuits. with excellent drift and noise characteristics, vref offers a good voltage reference for data acquisition and transducer excitation applications as well. output drift is typically better than ?0 ppm/ c, with 315 nv/ hz (typ) noise spectral density at 1 khz.
rev. d tmp01 C9C preserving accuracy over wide temperature range operation the tmp01 is un ique in offering both a wide range temperature sensor and the associated detection circuitry needed to imple- ment a complete thermostatic control function in one monolithic device. while the voltage reference, setpoint comparators, and output buffer amplifiers have been carefully compensated to maintain accuracy over the specified temperature range, the user has an additional task in maintaining the accuracy over wide operating temperature ranges in this application. since the tmp01 is both sensor and control circuit, in many applications, it is possible that the external components used to program and interface the device may be subjected to the same temperature extremes. thus, it may be necessary to locate components in close thermal proximity to minimize large temperature differen- tials, and to account for thermal drift errors such as resistor matching tempcos, amplifier error drift, and the like where appropriate. circuit design with the tmp01 requires a slightly different perspective regarding the thermal behavior of elec- tronic components. thermal response time the time required for a temperature sensor to settle to a speci- fied accuracy is a function of the thermal mass of the sensor, and the thermal conductivity between the sensor and the object being sensed. thermal mass is often considered equivalent to capacitance. thermal conductivity is commonly specified using the symbol q, and can be thought of as the reciprocal of thermal resistance. it is commonly specified in units of degrees per watt of power transferred across the thermal joint. thus, the time required for the tmp01 to settle to the desired accuracy is dependent on the package selected, the thermal contact estab- lished in that particular application, and the equivalent power of the heat source. in most applications, the settling time is prob- ably best determined empirically. switching loads with the open-collector outputs in many temperature sensing and control applications, some type of switching is required. whether it be to turn on a heater when the temperature goes below a minimum value or to turn off a motor that is overheating, the open-collector outputs over and under can be used. for the majority of applications, the switches used need to handle large currents on the order of 1 a and above. because the tmp01 is accurately measuring tempera- ture, the open-collector outputs should handle less than 20 ma of current to minimize self-heating. clearly, the overtemp and undertemp outputs should not drive the equipment directly. instead, an external switching device is required to handle the large currents. some examples of these are relays, power mosfets, thyristors, igbts, and darlingtons. figures 5a through 5e show a variety of circuits where the tmp01 controls a switch. the main consideration in these circuits, such as the relay in figure 5a, is the current required to activate the switch. motor shutdown 2604-12-311 coto in4001 or equiv. 12v r1 r2 r3 temperature sensor and voltage reference 1 2 3 4 7 hysteresis generator window comparator tmp01 vptat vref 8 5 6 figure 6a. reed relay drive it is important to check the particular relay to ensure that the current needed to activate the coil does not exceed the tmp01? recommended output current of 20 ma. this is easily deter- mined by dividing the relay coil voltage by the specified coil resistance. keep in mind that the inductance of the relay will create large voltage spikes that can damage the tmp01 output unless protected by a commutation diode across the coil, as shown. the relay shown has a contact rating of 10 w maximum. if a relay capable of handling more power is desired, the larger con- tacts will probably require a commensurately larger coil, with lower coil resistance and thus higher trigger current. as the contact power handling capability increases, so does the current needed for the coil. in some cases, an external driving transistor should be used to remove the current load on the tmp01 as explained in the next section. power fets are popular for handling a variety of high current dc loads. figure 5b shows the tmp01 driving a p-channel mosfet transistor for a simple heater circuit. when the out- put transistor turns on, the gate of the mosfet is pulled down to approximately 0.6 v, turning it on. for most mosfets, a gate-to-source voltage or vgs on the order of ? v to ? v is sufficient to turn the device on. figure 5c shows a similar circuit for turning on an n-channel mosfet, except that now the gate to source voltage is positive. for this reason, an external transis- tor must be used as an inverter so that the mosfet will turn on when the ?nder temp?output pulls down. nc = no connect nc nc irfr9024 or equiv. heating element 2.4k (12v) 1.2k (6v) 5% v+ r1 r2 r3 temperature sensor and voltage reference 1 2 3 4 7 hysteresis generator window comparator tmp01 vptat vref 8 5 6 figure 6b. driving a p-channel mosfet
rev. d tmp01 C10C irf130 nc = no connect nc nc 2n1711 heating element v+ r1 r2 r3 4.7k 4.7k temperature sensor and voltage reference 1 2 3 4 hysteresis generator window comparator tmp01 vptat vref 7 8 5 6 figure 6c. driving a n-channel mosfet isolated gate bipolar transistors (igbt) combine many of the benefits of power mosfets with bipolar transistors, and are used for a variety of high power applications. because igbts have a gate similar to mosfets, turning on and off the devices is relatively simple as shown in figure 5d. the turn-on voltage for the igbt shown (irgbc40s) is between 3.0 v and 5.5 v. this part has a continuous collector current rating of 50 a and a maximum collector to emitter voltage of 600 v, enabling it to work in very demanding applications. irgbc40s nc = no connect nc nc 2n1711 v+ r1 r2 r3 4.7k 4.7k temperature sensor and voltage reference 1 2 3 4 hysteresis generator window comparator tmp01 vptat vref 7 8 5 6 motor control figure 6d. driving an igbt the last class of high power devices discussed here are thyristors, which includes scrs and triacs. triacs are a useful alternative to relays for switching ac line voltages. the 2n6073a shown in figure 5e is rated to handle 4a (rms). the optoisolated moc3011. triac shown features excellent electrical isolation from the noisy ac line and complete control over the high power triac with only a few additional components. nc = no connect nc nc v+ = 5v r1 r2 r3 300 150 temperature sensor and voltage reference 1 2 3 4 hysteresis generator window comparator tmp01 vptat vref 7 8 5 6 moc3011 1 2 34 5 6 load ac 2n6073a figure 6e. controlling the 2n6073a triac high current switching as mentioned above, internal dissipation due to large loads on the tmp01 outputs will cause some temperature error due to self-heating. external transistors remove the load from the tmp01, so that virtually no power is dissipated in the internal transistors and no self-heating occurs. figures 6a through 6c show a few examples using external transistors. the simplest case, using a single transistor on the output to invert the output signal is shown in figure 6a. when the open collector of the tmp01 turns ?n and pulls the output down, the external transistor q1? base will be pulled low, turning off the transistor. another transistor can be added to reinvert the signal as shown in figure 6b. now, when the output of the tmp01 is pulled down, the first transis- tor, q1, turns off and its collector goes high, which turns q2 on, pulling its collector low. thus, the output taken from the collector of q2 is identical to the output of the tmp01. by picking a transistor that can accommodate large amounts of current, many high power devices can be switched. 2n1711 v+ r1 r2 r3 4.7k temperature sensor and voltage reference 1 2 3 4 hysteresis generator window comparator tmp01 vptat vref 7 8 5 6 i c q1 figure 7a. an external resistor minimizes self-heating q1 q2 2n1711 v+ r1 r2 r3 4.7k temperature sensor and voltage reference 1 2 3 4 hysteresis generator window comparator tmp01 vptat vref 7 8 5 6 i c 4.7k 2n1711 figure 7b. second transistor maintains polarity of tmp01 output an example of a higher power transistor is a standard darlington configuration as shown in figure 6c. the part chosen, tip-110, can handle 2 a continuous which is more than enough to con- trol many high power relays. in fact, the darlington itself can be used as the switch, similar to mosfets and igbts.
rev. d tmp01 C11C motor switch relay 12v 2n1711 v+ r1 r2 r3 4.7k temperature sensor and voltage reference 1 2 3 4 hysteresis generator window comparator tmp01 vptat vref 7 8 5 6 i c 4.7k tip-110 figure 7c. darlington transistor can handle large currents buffering the temperature output pin the vptat sensor output is a low impedance dc output voltage with a 5 mv/k temperature coefficient, and is useful in a number of measurement and control applications. in many applications, this voltage needs to be transmitted to a central location for processing. the buffered vptat voltage output is capable of 500 a drive into 50 pf (max). as mentioned in the discussion above regarding buffering circuits for the vref output, it is useful to consider external amplifiers for interfacing vptat to external circuitry to ensure accuracy, and to minimize loading which could create dissipation-induced temperature sensing errors. an excellent general purpose buffer circuit using the op177 is shown in figure 7. it is capable of driving over 10 ma, and will remain stable under capacitive loads of up to 0.1 f. other interfacing ideas are shown below. differential transmitter in noisy industrial environments, it is difficult to send an accurate analog signal over a significant distance. however, by sending the signal differentially on a wire pair, these errors can be signifi- cantly reduced. since the noise will be picked up equally on both wires, a receiver with high common-mode input rejection can be used to cancel out the noise very effectively at the receiving 0.1 f v+ v c l v out vptat op177 v+ r1 r2 r3 100 temperature sensor and voltage reference 1 2 3 4 hysteresis generator window comparator tmp01 vptat vref 7 8 5 6 10k figure 8. buffer vptat to handle difficult loads end. figure 8 shows two amplifiers being used to send the signal differentially, and an excellent differential receiver, the amp03, which features a common-mode rejection ratio of 95 db at dc and very low input and drift errors. v out vptat 1/2 op297 v+ r1 r2 r3 50 temperature sensor and voltage reference 1 2 3 4 hysteresis generator window comparator tmp01 vptat vref 7 8 5 6 10k v+ v 10k 1/2 op297 10k amp03 50 figure 9. send the signal differentially for noise immunity
rev. d tmp01 C12C 4 ma to 20 ma current loop another very common method of transmitting a signal over long distances is to use a 4 ma to 20 ma loop, as shown in figure 9. an advantage of using a 4 ma to 20 ma loop is that the accuracy of a current loop is not compromised by voltage drops across the line. one requirement of 4 ma to 20 ma circuits is that the remote end must receive all of its power from the loop, mean- ing that the circuit must consume less than 4 ma. operating from 5 v, the quiescent current of the tmp01 is 500 a max, and the op90s is 20 a max, totaling less than 4 ma. although not shown, the open collector outputs and temperature setting pins can be connected to do any local control of switching. the current is proportional to the voltage on the vptat output, and is calibrated to 4 ma at a temperature of ?0 c, to 20 ma for +85 c. the main equation governing the operation of this circuit gives the current as a function of vptat: i out = 1 r 6 vptat r 5 r 2 vref r 3 r 3 + r 1 1 + r 5 r 2 ? ? ? ? ? ? ? ? ? ? ? ? the resulting temperature coefficient of the output current is 128 a/ c. 5 8 1 4 r l 2n1711 vref gnd v+ vptat tmp01 r5 100k r2 39.2k 7 6 4 3 2 op90 r1 243k r3 100k r6 100 4 20ma 5v to 13.2v figure 10. 4ma to 20 ma current loop to determine the resistor values in this circuit, first note that vref remains constant over temperature. thus the ratio of r5 over r2 must give a variation of i out from 4 ma to 20 ma as vptat varies from 1.165 v at 40 c to 1.79 v at +85 c. the absolute value of the resistors is not important, only the ratio. for convenience, 100 k ? is chosen for r5. once r2 is calcu- lated, the value of r3 and r1 is determined by substituting 4 ma for i out and 1.165 v for vptat and solving. the final values are shown in the circuit. the op90 is chosen for this circuit because of its ability to operate on a single supply and its high accuracy. for initial accuracy, a 10 k ? trim potentiometer can be included in series with r3, and the value of r3 lowered to 95 k ? . the potentiometer should be adjusted to produce an output current of 12.3 ma at 25 c. temperature-to-frequency converter another common method of transmitting analog information is to convert a voltage to the frequency domain. this is easily done with any of the low cost monolithic voltage-to-frequency converters (vfcs) available, which feature a robust, open- collector digital output. a digital signal is very immune to noise and voltage drops because the only important information is the frequency. as long as the conversions between temperature and frequency are done accurately, the temperature data can be successfully transmitted. a simple circuit to do this combines the tmp01 with an ad654 vfc, as shown in figure 10. the ad654 outputs a square wave that is proportional to the dc input voltage according to the following equation: f out = v in 10 ( r 1 + r 2) c t by simply connecting the vptat output to the input of the ad654, the 5 mv/ c temperature coefficient gives a sensitivity of 25 hz/ c, centered around 7.5 khz at 25 c. the trimming resistor r2 is needed to calibrate the absolute accuracy of the ad654. for more information on that part, consult the ad654 data sheet. finally, the ad650 can be used to accurately convert the frequency back to a dc voltage on the receiving end. 4 3 7 6 8 1 2 5 ad654 vptat v+ r1 r2 r3 temperature sensor and voltage reference 1 2 3 4 hysteresis generator window comparator tmp01 vptat vref 7 8 5 6 r1 1.8k osc v+ f out c t 0.1 f 5k v+ r2 500 figure 11. temperature-to-frequency converter
rev. d tmp01 C13C in4148 i 1 i 2 6 5 3 4 1 2 2.5v v+ ref43 4 6 2 1.16v to 1.7v isolation barrier op290 v+ v+ r1 r2 r3 100 temperature sensor and voltage reference 1 2 3 4 hysteresis generator window comparator tmp01 vptat vref 7 8 5 6 7 6 4 3 2 op290 680pf r1 470k v+ 7 6 4 3 2 op90 604k 100k 680pf il300xc figure 12. isolation amplifier isolation amplifier in many industrial applications, the sensor is located in an envi- ronment that needs to be electrically isolated from the central processing area. figure 11 shows a simple circuit that uses an 8-pin optoisolator (il300xc) that can operate across a 5,000 v barrier. ic1 (an op290 single-supply amplifier) is used to drive the led connected between pins 1 to 2. the feedback actually comes from the photodiode connected from pins 3 to 4. the op290 drives the led such that there is enough current gener- ated in the photodiode to exactly equal the current derived from the vptat voltage across the 470 k ? resistor. on the receiving end, an op90 converts the current from the second photodiode to a voltage through its feedback resistor r2. note that the other amplifier in the dual op290 is used to buffer the 2.5 v reference voltage of the tmp01 for an accurate, low drift led bias level without affecting the programmed hysteresis current. a ref43 (a precision 2.5 v reference) provides an accurate bias level at the receiving end. to understand this circuit, it helps to examine the overall equation for the output voltage. first, the current (i1) in the photodiode is set by: i v vptat k 1 25 470 = ? . ? note that the il300xc has a gain of 0.73 (typical) with a min and max of 0.693 and 0.769 respectively. since this is less than 1.0, r2 must be larger than r1 to achieve overall unity gain. to show this, the full equation is: vvirv v vptat k k vptat out =?=? ? ? ? ? ? ? ? = 25 25 07 25 470 644 22 ... . ? ? a trim is included for r2 to correct for the initial gain accuracy of the il300xc. to perform this trim, simply adjust for an output voltage equal to vptat at any particular temperature. for example, at room temperature, vptat = 1.49 v, so adjust r2 until v out = 1.49 v as well. both the ref43 and the op90 operate from a single supply, and contribute no significant error due to drift. in order to avoid the accuracy trim, and to reduce board space, complete isolation amplifiers are available, such as the high accuracy ad202. out-of-range warning by connecting the two open collector outputs of the tmp01 together into a wired-or configuration, a temperature out- of-range warning signal is generated. this can be useful in sensitive equipment calibrated to work over a limited temperature range. r1, r2, and r3 in figure 11 are chosen to give a tempera- ture range of 10 c around room temperature (25 c). thus, if the temperature in the equipment falls below 15 c or rises above 35 c, the undertemp output or overtemp output respectively will go low and turn the led on. the led may be replaced with a simple pull-up resistor to give a logic output for control- ling the instrument, or any of the switching devices discussed above can be used. led vptat v+ r1 47.5k r2 4.99k r3 71.5k 200 temperature sensor and voltage reference 1 2 3 4 hysteresis generator window comparator tmp01 vptat vref 7 8 5 6 figure 13. out-of-range warning
rev. d tmp01 C14C translating 5 mv/k to 10 mv/ c a useful c ircuit shown in figure 13 tran slates the vptat output voltage, which is calibrated in kelvins, into an output that can be read directly in degrees celsius on a voltmeter display. to accomplish this, an external amplifier is configured as a differen- tial amplifier. the resistors are scaled so the vref voltage will exactly cancel the vptat voltage at 0.0 c. 5 1 +15v 15v 10pf v out (10mv/ c) (v out = 0.0v @ t = 0.0 c) 487 7 6 4 3 2 op177 100k 100k 4.12k vptat vref tmp01 4.22k 105k figure 14. translating 5 mv/k to 10 mv/ c however, the gain from vptat to the output is two, so that 5 mv/k becomes 10 mv/ c. thus, for a temperature of 80 c, the output voltage is 800 mv. circuit errors will be due prima- rily to the inaccuracies of the resistor values. using 1% resistors, the observed error was less than 10 mv, or 1 c. the 10 pf feedback capacitor helps to ensure against oscillations. for better accuracy, a adjustment potentiometer can be added in series with either 100 k ? resistor. translating vptat to the fahrenheit scale a very similar circuit to the one shown in figure 13 can be used to translate vptat into an output that can be read directly in degrees fahrenheit, with a scaling of 10 mv/ f. only unity gain or less is available from the first stage differentiating circuit, so the second amplifier provides a gain of two to complete the conversion to the fahrenheit scale. using the circuit in figure 14, a temperature of 0.0 f gives an output of 0.00 v. at room temperature (70 f), the output voltage is 700 mv. a 40 c to +85 c operating range translates into 40 f to +185 f. the errors are essentially the same as for the circuit in figure 13. v out = 0.0v @ t = 0.0 f (10mv/ f) 5 1 +15v 15v 10pf 121 7 6 4 3 2 1/2 op297 100k 100k 6.49k vptat vref tmp01 1.0k 90.9k 100k 7 6 5 1/2 op297 100k figure 15. translating 5 mv/k to 10 mv/ f
rev. d tmp01 C15C outline dimensions dimensions shown in inches and (mm). 8-lead epoxy dip (n-8) 0.160 (4.06) 0.115 (2.93) 0.130 (3.30) min 0.210 (5.33) max 0.015 (0.381) typ 0.430 (10.92) 0.348 (8.84) 0.280 (7.11) 0.240 (6.10) 4 5 8 1 0.070 (1.77) 0.045 (1.15) 0.022 (0.558) 0.014 (0.356) 0.325 (8.25) 0.300 (7.62) 0 - 15 0.100 (2.54) bsc 0.015 (0.381) 0.008 (0.204) seating plane 0.195 (4.95) 0.115 (2.93) 8-lead soic (r-8) seating plane 0.0500 (1.27) bsc 4 5 8 1 0.2440 (6.20) 0.2284 (5.80) 0.1574 (4.00) 0.1497 (3.80) 0.1968 (5.00) 0.1890 (4.80) 0.0500 (1.27) 0.0160 (0.41) 0 -8 45 0.0196 (0.50) 0.0099 (0.25) 0.0098 (0.25) 0.0075 (0.19) 0.102 (2.59) 0.094 (2.39) 0.0192 (0.49) 0.0138 (0.35) 0.0098 (0.25) 0.0040 (0.10) 8-lead to-99 (to-99) 45 bsc 0.115 (2.92) bsc 0.034 (0.86) 0.027 (0.69) 0.045 (1.14) 0.027 (0.69) 0.160 (4.06) 0.110 (2.79) 0.115 (2.92) bsc 0.230 (5.84) bsc 6 8 5 7 1 4 2 3 reference plane base & seating plane 0.335 (8.51) 0.305 (7.75) 0.370 (9.40) 0.335 (8.51) 0.750 (19.05) 0.500 (12.70) 0.045 (1.14) 0.010 (0.25) 0.050 (1.27) max 0.040 (1.02) max 0.019 (0.48) 0.016 (0.41) 0.021 (0.53) 0.016 (0.41) 0.185 (4.70) 0.165 (4.19) 0.250 (6.35) min
rev. d tmp01 C16C c00333C0C3/02 (d) printed in u.s.a. revision history location page 1/02?ata sheet changed from rev. c to rev. d. edits to general description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 edits to specifications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 edits to wafer test limits . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 edits to dice characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 edits to ordering guide . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5

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